Apparatus for establishing signal coupling between a signal line and an antenna structure

ABSTRACT

An apparatus for coupling a signal supply with an antenna including first and second elements. The signal supply delivers a signal to the antenna at a connection locus. The first element has a first edge and the second element has a second edge; the connection locus includes part of the first and second edges. The apparatus includes a first and second feed structure. The first feed structure extends a feed distance from the signal supply to the second edge and divides the first element into two lands in spaced relation with the first feed structure to establish a separation distance intermediate the first feed structure and the two lands. The second feed structure couples the signal supply with the first proximal edge. The separation distance establishes a signal transmission structure between the two lands and the first feed structure.

[0001] This is a divisional of application U.S. Ser. No. 09/855,413,filed May 15, 2001. (Attorney Docket No. DDM01-001)

BACKGROUND OF THE INVENTION

[0002] 1. Field of the Invention

[0003] The present invention relates generally to electromagnetic energyradiation and reception, and especially relates to electromagneticenergy radiation and reception effected using impulse radio energy.Still more particularly the present invention provides an antenna withan adjustable-impedance feed that is suited for broadband energyradiation and reception, and particularly well suited for broadbandenergy radiation and reception employing impulse radio energy.

[0004] 2. Related Art

[0005] Recent advances in communications technology have enabled anemerging, revolutionary ultra wideband technology (UWB) called impulseradio communications systems (hereinafter called impulse radio).

[0006] Impulse radio was first fully described in a series of patents,including U.S. Pat. Nos. 4,641,317 (issued Feb. 3, 1987), 4,813,057(issued Mar. 14, 1989), 4,979,186 (issued Dec. 18, 1990) and 5,363,108(issued Nov. 8, 1994) to Larry W. Fullerton. A second generation ofimpulse radio patents include U.S. Pat. Nos. 5,677,927 (issued Oct. 14,1997) to Fullerton et al; and 5,687,169 (issued Nov. 11, 1997) and5,832,035 (issued Nov. 3, 1998) to Fullerton. These patent documents areincorporated herein by reference.

[0007] Uses of impulse radio systems are described in U.S. patentapplication Ser. No. 09/332,502, entitled, “System and Method forIntrusion Detection Using a Time Domain Radar Array,” and U.S. patentapplication Ser. No. 09/332,503, entitled, “Wide Area Time Domain RadarArray,” both filed Jun. 14, 1999, both of which are assigned to theassignee of the present invention, and both of which are incorporatedherein by reference.

[0008] Earlier-filed applications relating to impulse radio antenna artsinclude U.S. patent application Ser. No. 09/652,282, entitled,“Semi-Coaxial Horn”, filed Aug. 30, 2000; U.S. patent application Ser.No. 09/670,972, entitled, “Electromagnetic Antenna Apparatus”, filedSep. 27, 2000; U.S. patent application Ser. No. 09/753,243, entitled,“Planar Loop Antenna”, filed Jan. 2, 2001; and U.S. patent applicationSer. No. 09/753,244, entitled, “Single Element Antenna Apparatus”, filedJan. 2, 2001.

[0009] Basic impulse radio transmitters emit short pulses approaching aGaussian monocycle with tightly controlled pulse-to-pulse intervals.Impulse radio systems typically use pulse position modulation, which isa form of time modulation where the value of each instantaneous sampleof a modulating signal is caused to modulate the position of a pulse intime.

[0010] For impulse radio communications, the pulse-to-pulse interval isvaried on a pulse-by-pulse basis by two components: an informationcomponent and a pseudo-random code component. Unlike direct sequencespread spectrum systems, the pseudo-random code for impulse radiocommunications is not necessary for energy spreading because themonocycle pulses themselves have an inherently wide bandwidth. Instead,the pseudo-random code of an impulse radio system is used forchannelization, energy smoothing in the frequency domain and forinterference suppression.

[0011] Generally speaking, an impulse radio receiver is a directconversion receiver with a cross correlator front end. The front endcoherently converts an electromagnetic pulse train of monocycle pulsesto a baseband signal in a single stage. The data rate of the impulseradio transmission is typically a fraction of the periodic timing signalused as a time base. Because each data bit modulates the time positionof many pulses of the periodic timing signal, this yields a modulated,coded timing signal that comprises a train of identically shaped pulsesfor each single data bit. The impulse radio receiver integrates multiplepulses to recover the transmitted information.

[0012] Antennas having ultra-wide band (UWB) properties are desired fora variety of applications, including impulse radio applications forcommunications, positioning, and other uses. Historically the principaluse of UWB antennas has been in multi-band communication systems. Suchmulti-band communication systems require an ultra-wide band antenna thatcan handle narrow band signals at a variety of frequencies.

[0013] The recently emerging impulse radio communications technologyoften referred to as impulse radio has placed different, more stringentrequirements on antenna performance. Impulse radio communications usesUWB signals, so an antenna for use in an impulse radio system musttransmit or receive (or, transmit and receive) over all frequenciesacross an ultra-wide band at the same time. Thus, ultra-wide bandimpulse radio requires that an antenna performs well over ultra-widebandwidths, but is also non-dispersive of those signals. It is desirablein such UWB impulse radio systems to have an antenna with a phase centerthat remains fixed as a function of frequency so that radiated andreceived waveforms are not distorted.

[0014] Many of the known UWB antennas do not meet this requirement. Themost frequently used class of UWB antennas is a “self similar” frequencyindependent antenna, such as a log periodic antenna or a spiral antenna.Such antennas rely on a smaller scale portion to radiate higherfrequency components, and a larger scale portion to radiate lowerfrequency components. As a result, different frequency components areradiated from different parts of the antenna, and resulting radiatedwaveforms are distorted. The distortion thus created can be correctedand compensated for by a variety of techniques known to artisans skilledin radio frequency (RF) design and signal processing. However, suchcorrective measures and structures add unnecessary complications andexpense to overall system design.

[0015] Horn-type antennas can be non-dispersive, but they tend to belarge, bulky and highly directive. Small element antennas are known,such as bow-tie antennas, but they tend to have excessive reflectionsthat can be offset only by resistive loading. Resistive loading is alossy solution that minimizes reflection at the cost of loweringradiation efficiency. Non-resistive loaded small element antennas havebeen disclosed in U.S. Pat. No. 5,363,108 (issued Nov. 8, 1994) to LarryW. Fullerton. These antennas emit a short non-dispersive pulse, but tendto have significant reflections and less than desirable impedancematching.

[0016] There is a need for a small omni-directional antenna that canradiate energy efficiently with minimal reflection and distortion.

[0017] In particular, there is a need for a small omni-directionalplanar dipole antenna that can radiate energy efficiently with minimalreflection and distortion.

SUMMARY OF THE INVENTION

[0018] An apparatus for establishing signal coupling between a signalsupply and an antenna structure that includes a first radiating elementand a second radiating element. The signal supply delivers a signal tothe antenna structure at a connection locus. The first radiating elementhas a first proximal edge and a first distal edge with respect to thesignal supply in an installed orientation. The second radiating elementhas a second proximal edge and a second distal edge with respect to thesignal supply in the installed orientation. The connection locusgenerally includes a portion of the first proximal edge and the secondproximal edge. The apparatus includes: (a) a first feed structureextending a feed distance from the signal supply in the installedorientation to the second proximal edge. The first feed structuresubstantially divides the first radiating element into at least twoelectrically common lands in spaced relation with the first feedstructure to establish a separation distance intermediate the first feedstructure and the at least two lands on two sides of the first feedstructure substantially along the feed distance. (b) a second feedstructure coupling the signal supply with the first proximal edge. Theseparation distance is dimensioned appropriately to establish a signaltransmission structure between the at least two lands and the first feedstructure. The geometry of the signal transmission structure may bevaried along its length so as to establish a desired impedancetransformation intermediate the first and second feed structures.

[0019] It is therefore an object of the present invention to provide asmall omni-directional antenna that can radiate energy efficiently withminimal reflection and distortion.

[0020] It is a further object of the present invention to provide asmall omni-directional planar dipole antenna that can radiate energyefficiently with minimal reflection and distortion.

[0021] Further objects and features of the present invention will beapparent from the following specification and claims when considered inconnection with the accompanying drawings, in which like elements arelabeled using like reference numerals in the various figures,illustrating the preferred embodiments of the invention.

BRIEF DESCRIPTION OF THE DRAWINGS

[0022]FIG. 1A illustrates a representative Gaussian Monocycle waveformin the time domain.

[0023]FIG. 1B illustrates the frequency domain amplitude of the GaussianMonocycle of FIG. 1A.

[0024]FIG. 2A illustrates a pulse train comprising pulses as in FIG. 1A.

[0025]FIG. 2B illustrates the frequency domain amplitude of the waveformof FIG. 2A.

[0026]FIG. 3 illustrates the frequency domain amplitude of a sequence oftime coded pulses.

[0027]FIG. 4 illustrates a typical received signal and interferencesignal.

[0028]FIG. 5A illustrates a typical geometrical configuration givingrise to multipath received signals.

[0029]FIG. 5B illustrates exemplary multipath signals in the timedomain.

[0030] FIGS. 5C-5E illustrate a signal plot of various multipathenvironments.

[0031]FIG. 5F illustrates the Rayleigh fading curve associated withnon-impulse radio transmissions in a multipath environment.

[0032]FIG. 5G illustrates a plurality of multipaths with a plurality ofreflectors from a transmitter to a receiver.

[0033]FIG. 5H graphically represents signal strength as volts vs. timein a direct path and multipath environment.

[0034]FIG. 6 illustrates a representative impulse radio transmitterfunctional diagram.

[0035]FIG. 7 illustrates a representative impulse radio receiverfunctional diagram.

[0036]FIG. 8A illustrates a representative received pulse signal at theinput to the correlator.

[0037]FIG. 8B illustrates a sequence of representative impulse signalsin the correlation process.

[0038]FIG. 8C illustrates the output of the correlator for each of thetime offsets of FIG. 8B.

[0039] FIGS. 9(A) through (D) are schematic illustrations ofrepresentative planar dipole antennas with elliptical elements havingvarious ratios of major-to-minor axes.

[0040]FIG. 10 is a schematic diagram of detail of an antenna feedstructure for a spheroidal dipole antenna.

[0041]FIG. 11(A) is a side view of a right angle coaxial connector feedstructure with a planar antenna.

[0042]FIG. 11(B) is a side view of a straight coaxial connector feedstructure with a planar antenna.

[0043]FIG. 11(C) is a top view of a curved feed interface arrangementfor an antenna of the sort illustrated in FIG. 11(A) or FIG. 11(B) takenalong Section 11C-11C of FIG. 11(A) or FIG. 11(B).

[0044]FIG. 12 is a schematic plan view of a planar antenna configuredaccording to the construction illustrated in FIG. 11.

[0045]FIG. 13 is a schematic plan view of a planar dipole antennaconfigured according to the preferred embodiment of the presentinvention.

[0046]FIG. 14 is a schematic perspective view of a spheroidal dipoleantenna configured according to the present invention.

[0047]FIG. 15 is a graphic representation of a typical relationshipbetween antenna gain and signal frequency for antennas configuredaccording to the teachings of the present invention, includingadjustment of gain roll-off for such antennas.

[0048]FIG. 16 is a schematic plan view of a planar dipole antennaconfigured according to an alternate embodiment of the presentinvention.

DETAILED DESCRIPTION OF THE EMBODIMENTS

[0049] Overview of the Invention

[0050] The present invention will now be described more fully in detailwith reference to the accompanying drawings, in which the preferredembodiments of the invention are shown. This invention should not,however, be construed as limited to the embodiments set forth herein;rather, they are provided so that this disclosure will be thorough andcomplete and will fully convey the scope of the invention to thoseskilled in art. Like numbers refer to like elements throughout.

[0051] Impulse Radio Basics

[0052] This section is directed to technology basics and provides thereader with an introduction to impulse radio concepts, as well as otherrelevant aspects of communications theory. This section includessubsections relating to waveforms, pulse trains, coding for energysmoothing and channelization, modulation, reception and demodulation,interference resistance, processing gain, capacity, multipath andpropagation, distance measurement, and qualitative and quantitativecharacteristics of these concepts. It should be understood that thissection is provided to assist the reader with understanding the presentinvention, and should not be used to limit the scope of the presentinvention.

[0053] Impulse radio refers to a radio system based on short, low dutycycle pulses. An ideal impulse radio waveform is a short Gaussianmonocycle. As the name suggests, this waveform attempts to approach onecycle of radio frequency (RF) energy at a desired center frequency. Dueto implementation and other spectral limitations, this waveform may bealtered significantly in practice for a given application. Mostwaveforms with enough bandwidth approximate a Gaussian shape to a usefuldegree.

[0054] Impulse radio can use many types of modulation, including AM,time shift (also referred to as pulse position) and M-ary versions. Thetime shift method has simplicity and power output advantages that makeit desirable. In this document, the time shift method is used as anillustrative example.

[0055] In impulse radio communications, the pulse-to-pulse interval canbe varied on a pulse-by-pulse basis by two components: an informationcomponent and a pseudo-random code component. Generally, conventionalspread spectrum systems make use of pseudo-random codes to spread thenormally narrow band information signal over a relatively wide band offrequencies. A conventional spread spectrum receiver correlates thesesignals to retrieve the original information signal. Unlike conventionalspread spectrum systems, the pseudo-random code for impulse radiocommunications is not necessary for energy spreading because themonocycle pulses themselves have an inherently wide bandwidth. Instead,the pseudo-random code is used for channelization, energy smoothing inthe frequency domain, resistance to interference, and reducing theinterference potential to nearby receivers.

[0056] The impulse radio receiver is typically a direct conversionreceiver with a cross correlator front end in which the front endcoherently converts an electromagnetic pulse train of monocycle pulsesto a baseband signal in a single stage. The baseband signal is the basicinformation signal for the impulse radio communications system. It isoften found desirable to include a subcarrier with the baseband signalto help reduce the effects of amplifier drift and low frequency noise.The subcarrier that is typically implemented alternately reversesmodulation according to a known pattern at a rate faster than the datarate. This same pattern is used to reverse the process and restore theoriginal data pattern just before detection. This method permitsalternating current (AC) coupling of stages, or equivalent signalprocessing to eliminate direct current (DC) drift and errors from thedetection process. This method is described in detail in U.S. Pat. No.5,677,927 to Fullerton et al.

[0057] In impulse radio communications utilizing time shift modulation,each data bit typically time position modulates many pulses of theperiodic timing signal. This yields a modulated, coded timing signalthat comprises a train of identically shaped pulses for each single databit. The impulse radio receiver integrates multiple pulses to recoverthe transmitted information.

[0058] Waveforms

[0059] Impulse radio refers to a radio system based on short, low dutycycle pulses. In the widest bandwidth embodiment, the resulting waveformapproaches one cycle per pulse at the center frequency. In more narrowband embodiments, each pulse consists of a burst of cycles usually withsome spectral shaping to control the bandwidth to meet desiredproperties such as out of band emissions or in-band spectral flatness,or time domain peak power or burst off time attenuation.

[0060] For system analysis purposes, it is convenient to model thedesired waveform in an ideal sense to provide insight into the optimumbehavior for detail design guidance. One such waveform model that hasbeen useful is the Gaussian monocycle as shown in FIG. 1A. This waveformis representative of the transmitted pulse produced by a step functioninto an ultra-wideband antenna. The basic equation normalized to a peakvalue of 1 is as follows:${f_{mono}(t)} = {\sqrt{e}\left( \frac{t}{\sigma} \right)^{\frac{- t^{2}}{2\sigma^{2}}}}$

[0061] Where,

[0062] σ is a time scaling parameter,

[0063] t is time,

[0064] f_(mono)(t) is the waveform voltage, and

[0065] e is the natural logarithm base.

[0066] The frequency domain spectrum of the above waveform is shown inFIG. 1B. The corresponding equation is:${F_{mono}(f)} = {\left( {2\pi} \right)^{\frac{3}{2}}\sigma \quad f\quad ^{{- 2}{({{\pi\sigma}\quad f})}^{2}}}$

[0067] The center frequency (f_(c)) or frequency of peak spectraldensity is: $f_{c} = \frac{1}{2{\pi\sigma}}$

[0068] These pulses, or bursts of cycles, may be produced by methodsdescribed in the patents referenced above or by other methods that areknown to one of ordinary skill in the art. Any practical implementationwill deviate from the ideal mathematical model by some amount. In fact,this deviation from ideal may be substantial and yet yield a system withacceptable performance. This is especially true for microwaveimplementations, where precise waveform shaping is difficult to achieve.These mathematical models are provided as an aid to describing idealoperation and are not intended to limit the invention. In fact, anyburst of cycles that adequately fills a given bandwidth and has anadequate on-off attenuation ratio for a given application will serve thepurpose of this invention.

[0069] A Pulse Train

[0070] Impulse radio systems can deliver one or more data bits perpulse; however, impulse radio systems more typically use pulse trains,not single pulses, for each data bit. As described in detail in thefollowing example system, the impulse radio transmitter produces andoutputs a train of pulses for each bit of information.

[0071] Prototypes built by the inventors have pulse repetitionfrequencies including 0.7 and 10 megapulses per second (Mpps, where eachmegapulse is 10⁶ pulses). FIGS. 2A and 2B are illustrations of theoutput of a typical 10 Mpps system with uncoded, unmodulated, 0.5nanosecond (ns) pulses 102. FIG. 2A shows a time domain representationof this sequence of pulses 102. FIG. 2B, which shows 60 MHz at thecenter of the spectrum for the waveform of FIG. 2A, illustrates that theresult of the pulse train in the frequency domain is to produce aspectrum comprising a set of lines 204 spaced at the frequency of the 10Mpps pulse repetition rate. When the full spectrum is shown, theenvelope of the line spectrum follows the curve of the single pulsespectrum 104 of FIG. 1B. For this simple uncoded case, the power of thepulse train is spread among roughly two hundred comb lines. Each combline thus has a small fraction of the total power and presents much lessof an interference problem to receiver sharing the band.

[0072] It can also be observed from FIG. 2A that impulse radio systemstypically have very low average duty cycles resulting in average powersignificantly lower than peak power. The duty cycle of the signal in thepresent example is 0.5%, based on a 0.5 ns pulse in a 100 ns interval.

[0073] Coding for Energy Smoothing and Channelization

[0074] For high pulse rate systems, it may be necessary to more finelyspread the spectrum than is achieved by producing comb lines. This maybe done by pseudo-randomly positioning each pulse relative to itsnominal position.

[0075]FIG. 3 is a plot illustrating the impact of a pseudo-noise (PN)code dither on energy distribution in the frequency domain (Apseudo-noise, or PN code is a set of time positions defining thepseudo-random positioning for each pulse in a sequence of pulses). FIG.3, when compared to FIG. 2B, shows that the impact of using a PN code isto destroy the comb line structure and spread the energy more uniformly.This structure typically has slight variations which are characteristicof the specific code used.

[0076] The PN code also provides a method of establishing independentcommunication channels using impulse radio. PN codes can be designed tohave low cross correlation such that a pulse train using one code willseldom collide on more than one or two pulse positions with a pulsestrain using another code during any one data bit time. Since a data bitmay comprise hundreds of pulses, this represents a substantialattenuation of the unwanted channel.

[0077] Modulation

[0078] Any aspect of the waveform can be modulated to conveyinformation. Amplitude modulation, phase modulation, frequencymodulation, time shift modulation and M-ary versions of these have beenproposed. Both analog and digital forms have been implemented. Of these,digital time shift modulation has been demonstrated to have variousadvantages and can be easily implemented using a correlation receiverarchitecture.

[0079] Digital time shift modulation can be implemented by shifting thecoded time position by an additional amount (that is, in addition to PNcode dither) in response to the information signal. This amount istypically very small relative to the PN code shift. In a 10 Mpps systemwith a center frequency of 2 GHz., for example, the PN code may commandpulse position variations over a range of 100 ns; whereas, theinformation modulation may only deviate the pulse position by 150 ps.

[0080] Thus, in a pulse train of n pulses, each pulse is delayed adifferent amount from its respective time base clock position by anindividual code delay amount plus a modulation amount, where n is thenumber of pulses associated with a given data symbol digital bit.

[0081] Modulation further smooths the spectrum, minimizing structure inthe resulting spectrum.

[0082] Reception and Demodulation

[0083] Clearly, if there were a large number of impulse radio userswithin a confined area, there might be mutual interference. Further,while the PN coding minimizes that interference, as the number of usersrises, the probability of an individual pulse from one user's sequencebeing received simultaneously with a pulse from another user's sequenceincreases. Impulse radios are able to perform in these environments, inpart, because they do not depend on receiving every pulse. The impulseradio receiver performs a correlating, synchronous receiving function(at the RF level) that uses a statistical sampling and combining of manypulses to recover the transmitted information.

[0084] Impulse radio receivers typically integrate from 1 to 1000 ormore pulses to yield the demodulated output. The optimal number ofpulses over which the receiver integrates is dependent on a number ofvariables, including pulse rate, bit rate, interference levels, andrange.

[0085] Interference Resistance

[0086] Besides channelization and energy smoothing, the PN coding alsomakes impulse radios highly resistant to interference from all radiocommunications systems, including other impulse radio transmitters. Thisis critical as any other signals within the band occupied by an impulsesignal potentially interfere with the impulse radio. Since there arecurrently no unallocated bands available for impulse systems, they mustshare spectrum with other conventional radio systems without beingadversely affected. The PN code helps impulse systems discriminatebetween the intended impulse transmission and interfering transmissionsfrom others.

[0087]FIG. 4 illustrates the result of a narrow band sinusoidalinterference signal 402 overlaying an impulse radio signal 404. At theimpulse radio receiver, the input to the cross correlation would includethe narrow band signal 402, as well as the received ultrawide-bandimpulse radio signal 404. The input is sampled by the cross correlatorwith a PN dithered template signal 406. Without PN coding, the crosscorrelation would sample the interfering signal 402 with such regularitythat the interfering signals could cause significant interference to theimpulse radio receiver. However, when the transmitted impulse signal isencoded with the PN code dither (and the impulse radio receiver templatesignal 406 is synchronized with that identical PN code dither) thecorrelation samples the interfering signals pseudo-randomly. The samplesfrom the interfering signal add incoherently, increasing roughlyaccording to square root of the number of samples integrated; whereas,the impulse radio samples add coherently, increasing directly accordingto the number of samples integrated. Thus, integrating over many pulsesovercomes the impact of interference.

[0088] Processing Gain

[0089] Impulse radio is resistant to interference because of its largeprocessing gain. For typical spread spectrum systems, the definition ofprocessing gain, which quantifies the decrease in channel interferencewhen wide-band communications are used, is the ratio of the bandwidth ofthe channel to the bit rate of the information signal. For example, adirect sequence spread spectrum system with a 10 kHz informationbandwidth and a 10 MHz channel bandwidth yields a processing gain of1000 or 30 dB. However, far greater processing gains are achieved withimpulse radio systems, where for the same 10 kHz information bandwidthis spread across a much greater 2 GHz. channel bandwidth, thetheoretical processing gain is 200,000 or 53 dB.

[0090] Capacity

[0091] It has been shown theoretically, using signal to noise arguments,that thousands of simultaneous voice channels are available to animpulse radio system as a result of the exceptional processing gain,which is due to the exceptionally wide spreading bandwidth.

[0092] For a simplistic user distribution, with N interfering users ofequal power equidistant from the receiver, the total interference signalto noise ratio as a result of these other users can be described by thefollowing equation: $V_{tot}^{2} = \frac{N\quad \sigma^{2}}{\sqrt{Z}}$

[0093] Where V² _(tot) is the total interference signal to noise ratiovariance, at the receiver;

[0094] N is the number of interfering users;

[0095] σ² is the signal to noise ratio variance resulting from one ofthe interfering signals with a single pulse cross correlation; and

[0096] Z is the number of pulses over which the receiver integrates torecover the modulation.

[0097] This relationship suggests that link quality degrades graduallyas the number of simultaneous users increases. It also shows theadvantage of integration gain. The number of users that can be supportedat the same interference level increases by the square root of thenumber of pulses integrated.

[0098] Multipath and Propagation

[0099] One of the striking advantages of impulse radio is its resistanceto multipath fading effects. Conventional narrow band systems aresubject to multipath through the Rayleigh fading process, where thesignals from many delayed reflections combine at the receiver antennaaccording to their seemingly random relative phases. This results inpossible summation or possible cancellation, depending on the specificpropagation to a given location. This situation occurs where the directpath signal is weak relative to the multipath signals, which representsa major portion of the potential coverage of a radio system. In mobilesystems, this results in wild signal strength fluctuations as a functionof distance traveled, where the changing mix of multipath signalsresults in signal strength fluctuations for every few feet of travel.

[0100] Impulse radios, however, can be substantially resistant to theseeffects. Impulses arriving from delayed multipath reflections typicallyarrive outside of the correlation time and thus can be ignored. Thisprocess is described in detail with reference to FIGS. 5A and 5B. InFIG. 5A, three propagation paths are shown. The direct path representingthe straight line distance between the transmitter and receiver is theshortest. Path 1 represents a grazing multipath reflection, which isvery close to the direct path. Path 2 represents a distant multipathreflection. Also shown are elliptical (or, in space, ellipsoidal) tracesthat represent other possible locations for reflections with the sametime delay.

[0101]FIG. 5B represents a time domain plot of the received waveformfrom this multipath propagation configuration. This figure comprisesthree doublet pulses as shown in FIG. 1A. The direct path signal is thereference signal and represents the shortest propagation time. The path1 signal is delayed slightly and actually overlaps and enhances thesignal strength at this delay value. Note that the reflected waves arereversed in polarity. The path 2 signal is delayed sufficiently that thewaveform is completely separated from the direct path signal. If thecorrelator template signal is positioned at the direct path signal, thepath 2 signal will produce no response. It can be seen that only themultipath signals resulting from very close reflectors have any effecton the reception of the direct path signal. The multipath signalsdelayed less than one quarter wave (one quarter wave is about 1.5inches, or 3.5 cm at 2 GHz center frequency) are the only multipathsignals that can attenuate the direct path signal. This region isequivalent to the first Fresnel zone familiar to narrow band systemsdesigners. Impulse radio, however, has no further nulls in the higherFresnel zones. The ability to avoid the highly variable attenuation frommultipath gives impulse radio significant performance advantages.

[0102]FIG. 5A illustrates a typical multipath situation, such as in abuilding, where there are many reflectors 5A04, 5A05 and multiplepropagation paths 5A02, 5A01. In this figure, a transmitter TX 5A06transmits a signal which propagates along the multiple propagation paths5A02, 5A04 to receiver RX 5A08, where the multiple reflected signals arecombined at the antenna.

[0103]FIG. 5B illustrates a resulting typical received composite pulsewaveform resulting from the multiple reflections and multiplepropagation paths 5A01, 5A02. In this figure, the direct path signal5A01 is shown as the first pulse signal received. The multiple reflectedsignals (“multipath signals”, or “multipath”) comprise the remainingresponse as illustrated.

[0104]FIGS. 5C, 5D, and 5E represent the received signal from a TM-UWBtransmitter in three different multipath environments. These figures arenot actual signal plots, but are hand drawn plots approximating typicalsignal plots. FIG. 5C illustrates the received signal in a very lowmultipath environment. This may occur in a building where the receiverantenna is in the middle of a room and is one meter from thetransmitter. This may also represent signals received from somedistance, such as 100 meters, in an open field where there are noobjects to produce reflections. In this situation, the predominant pulseis the first received pulse and the multipath reflections are too weakto be significant. FIG. 5D illustrates an intermediate multipathenvironment. This approximates the response from one room to the next ina building. The amplitude of the direct path signal is less than in FIG.5C and several reflected signals are of significant amplitude. (Notethat the scale has been increased to normalize the plot.) FIG. 5Eapproximates the response in a severe multipath environment such as:propagation through many rooms; from comer to comer in a building;within a metal cargo hold of a ship; within a metal truck trailer; orwithin an intermodal shipping container. In this scenario, the main pathsignal is weaker than in FIG. 5D. (Note that the scale has beenincreased again to normalize the plot.) In this situation, the directpath signal power is small relative to the total signal power from thereflections.

[0105] An impulse radio receiver in accordance with the presentinvention can receive the signal and demodulate the information usingeither the direct path signal or any multipath signal peak havingsufficient signal to noise ratio. Thus, the impulse radio receiver canselect the strongest response from among the many arriving signals. Inorder for the signals to cancel and produce a null at a given location,dozens of reflections would have to be cancelled simultaneously andprecisely while blocking the direct path—a highly unlikely scenario.This time separation of multipath signals together with time resolutionand selection by the receiver permit a type of time diversity thatvirtually eliminates cancellation of the signal. In a multiplecorrelator rake receiver, performance is further improved by collectingthe signal power from multiple signal peaks for additional signal tonoise performance.

[0106] Where the system of FIG. 5A is a narrow band system and thedelays are small relative to the data bit time, the received signal is asum of a large number of sine waves of random amplitude and phase. Inthe idealized limit, the resulting envelope amplitude has been shown tofollow a Rayleigh probability distribution as follows:${p(r)} = {\frac{r}{\sigma^{2}}{\exp \left( \frac{- r^{2}}{2\sigma^{2}} \right)}}$

[0107] where r is the envelope amplitude of the combined multipathsignals, and

[0108] {square root}{square root over (2)} σ² is the RMS amplitude ofthe combined multipath signals.

[0109] This distribution shown in FIG. 5F. It can be seen in FIG. 5Fthat 10% of the time, the signal is more than 10 dB attenuated. Thissuggests that 10 dB fade margin is needed to provide 90% linkavailability. Values of fade margin from 10 to 40 dB have been suggestedfor various narrow band systems, depending on the required reliability.This characteristic has been the subject of much research and can bepartially improved by such techniques as antenna and frequencydiversity, but these techniques result in additional complexity andcost.

[0110] In a high multipath environment such as inside homes, offices,warehouses, automobiles, trailers, shipping containers, or outside inthe urban canyon or other situations where the propagation is such thatthe received signal is primarily scattered energy, impulse radio,according to the present invention, can avoid the Rayleigh fadingmechanism that limits performance of narrow band systems. This isillustrated in FIGS. 5G and 5H in a transmit and receive system in ahigh multipath environment 5G00, wherein the transmitter 5G06 transmitsto receiver 5G08 with the signals reflecting off reflectors 5G03 whichform multipaths 5G02. The direct path is illustrated as 5G01 with thesignal graphically illustrated at 5H02 with the vertical axis being thesignal strength in volts and horizontal axis representing time innanoseconds. Multipath signals are graphically illustrated at 5H04.

[0111] Distance Measurement and Position Location

[0112] Impulse systems can measure distances to extremely fineresolution because of the absence of ambiguous cycles in the waveform.Narrow band systems, on the other hand, are limited to the modulationenvelope and cannot easily distinguish precisely which RF cycle isassociated with each data bit because the cycle-to-cycle amplitudedifferences are so small they are masked by link or system noise. Sincethe impulse radio waveform has no multi-cycle ambiguity, this allowspositive determination of the waveform position to less than awavelength—potentially, down to the noise floor of the system. This timeposition measurement can be used to measure propagation delay todetermine link distance, and once link distance is known, to transfer atime reference to an equivalently high degree of precision. Theinventors of the present invention have built systems that have shownthe potential for centimeter distance resolution, which is equivalent toabout 30 ps of time transfer resolution. See, for example, commonlyowned, co-pending application Ser. No. 09/045,929, filed Mar. 23, 1998,titled “Ultrawide-Band Position Determination System and Method”, andSer. No. 09/083,993, filed May 26, 1998, titled “System and Method forDistance Measurement by Inphase and Quadrature Signals in a RadioSystem”, both of which are incorporated herein by reference. Finally,distance measuring and position location using impulse radio using aplurality of distance architectures is enabled in co-pending andcommonly owned U.S. patent application Ser. No. 09/456,409, filed Dec.8, 1999, titled, “System and Method for Person or Object PositionLocation Utilizing Impulse Radio.”

[0113] Exemplary Transceiver Implementation

[0114] Transmitter

[0115] An exemplary embodiment of an impulse radio transmitter 602 of animpulse radio communication system having one subcarrier channel willnow be described with reference to FIG. 6.

[0116] The transmitter 602 comprises a time base 604 that generates aperiodic timing signal 606. The time base 604 typically comprises avoltage controlled oscillator (VCO), or the like, having a high timingaccuracy and low jitter, on the order of picoseconds (ps). The voltagecontrol to adjust the VCO center frequency is set at calibration to thedesired center frequency used to define the transmitter's nominal pulserepetition rate. The periodic timing signal 606 is supplied to aprecision timing generator 608.

[0117] The precision timing generator 608 supplies synchronizing signals610 to the code source 612 and utilizes the code source output 614together with an internally generated subcarrier signal (which isoptional) and an information signal 616 to generate a modulated, codedtiming signal 618.

[0118] The code source 612 comprises a storage device such as a randomaccess memory (RAM), read only memory (ROM), or the like, for storingsuitable PN codes and for outputting the PN codes as a code signal 614.Alternatively, maximum length shift registers or other computationalmeans can be used to generate the PN codes.

[0119] An information source 620 supplies the information signal 616 tothe precision timing generator 608. The information signal 616 can beany type of intelligence, including digital bits representing voice,data, imagery, or the like, analog signals, or complex signals.

[0120] A pulse generator 622 uses the modulated, coded timing signal 618as a trigger to generate output pulses. The output pulses are sent to atransmit antenna 624 via a transmission line 626 coupled thereto. Theoutput pulses are converted into propagating electromagnetic pulses bythe transmit antenna 624. In the present embodiment, the electromagneticpulses are called the emitted signal, and propagate to an impulse radioreceiver 702, such as shown in FIG. 7, through a propagation medium,such as air, in a radio frequency embodiment. In a preferred embodiment,the emitted signal is wide-band or ultrawide-band, approaching amonocycle pulse as in FIG. 1A. However, the emitted signal can bespectrally modified by filtering of the pulses. This filtering willusually cause each monocycle pulse to have more zero crossings (morecycles) in the time domain. In this case, the impulse radio receiver canuse a similar waveform as the template signal in the cross correlatorfor efficient conversion.

[0121] Receiver

[0122] An exemplary embodiment of an impulse radio receiver 702(hereinafter called the receiver) for the impulse radio communicationsystem is now described with reference to FIG. 7. More specifically, thesystem illustrated in FIG. 7 is for reception of digital data whereinone or more pulses are transmitted for each data bit.

[0123] The receiver 702 comprises a receive antenna 704 for receiving apropagated impulse radio signal 706. A received signal 708 from thereceive antenna 704 is coupled to a cross correlator or sampler 710 toproduce a baseband output 712. The cross correlator or sampler 710includes multiply and integrate functions together with any necessaryfilters to optimize signal to noise ratio.

[0124] The receiver 702 also includes a precision timing generator 714,which receives a periodic timing signal 716 from a receiver time base718. This time base 718 is adjustable and controllable in time,frequency, or phase, as required by the lock loop in order to lock onthe received signal 708. The precision timing generator 714 providessynchronizing signals 720 to the code source 722 and receives a codecontrol signal 724 from the code source 722. The precision timinggenerator 714 utilizes the periodic timing signal 716 and code controlsignal 724 to produce a coded timing signal 726. The template generator728 is triggered by this coded timing signal 726 and produces a train oftemplate signal pulses 730 ideally having waveforms substantiallyequivalent to each pulse of the received signal 708. The code forreceiving a given signal is the same code utilized by the originatingtransmitter 602 to generate the propagated signal 706. Thus, the timingof the template pulse train 730 matches the timing of the receivedsignal pulse train 708, allowing the received signal 708 to besynchronously sampled in the correlator 710. The correlator 710 ideallycomprises a multiplier followed by a short term integrator to sum themultiplier product over the pulse interval. Further examples and detailsof correlation and sampling processes can be found in commonly ownedU.S. Pat. Nos. 4,641,317, 4,743,906, 4,813,057 and 4,979,186 which areincorporated herein by reference, and commonly owned and co-pendingapplication Ser No. 09/356,384, filed Jul. 16, 1999, titled: “BasebandSignal Converter Device for a Wideband Impulse Radio Receiver,” which isincorporated herein by reference.

[0125] The output of the correlator 710, also called a baseband signal712, is coupled to a subcarrier demodulator 732, which demodulates thesubcarrier information signal from the subcarrier. The purpose of theoptional subcarrier process, when used, is to move the informationsignal away from DC (zero frequency) to improve immunity to lowfrequency noise and offsets. The output of the subcarrier demodulator732 is then filtered or integrated in a pulse summation stage 734. Thepulse summation stage produces an output representative of the sum of anumber of pulse signals comprising a single data bit. The output of thepulse summation stage 734 is then compared with a nominal zero (orreference) signal output in a detector stage 738 to determine an outputsignal 739 representing an estimate of the original information signal616.

[0126] The baseband signal 712 is also input to a lowpass filter 742(also referred to as lock loop filter 742). A control loop comprisingthe lowpass filter 742, time base 718, precision timing generator 714,template generator 728, and correlator 710 is used to generate afiltered error signal 744. The filtered error signal 744 providesadjustments to the adjustable time base 718 to time position theperiodic timing signal 726 in relation to the position of the receivedsignal 708.

[0127] In a transceiver embodiment, substantial economy can be achievedby sharing part or all of several of the functions of the transmitter602 and receiver 702. Some of these include the time base 718, precisiontiming generator 714, code source 722, antenna 704, and the like.

[0128] FIGS. 8A-8C illustrate the cross correlation process and thecorrelation function. FIG. 8A shows the waveform of a template signal.FIG. 8B shows the waveform of a received impulse radio signal at a setof several possible time offsets. FIG. 8C represents the output of thecorrelator (multiplier and short time integrator) for each of the timeoffsets of FIG. 8B. Thus, this graph, FIG. 8C, does not show a waveformthat is a function of time, but rather a function of time-offset, i.e.,for any given pulse received, there is only one corresponding pointwhich is applicable on this graph. This is the point corresponding tothe time offset of the template signal used to receive that pulse.

[0129] Further examples and details of subcarrier processes andprecision timing can be found described in U.S. Pat. No. 5,677,927,titled “An Ultrawide-Band Communications System and Method”, andcommonly owned co-pending application Ser. No. 09/146,524, filed Sep. 3,1998, titled “Precision Timing Generator System and Method”, both ofwhich are incorporated herein by reference.

[0130] Impulse Radio as Used in the Present Invention

[0131] When utilized in a radio communication network, thecharacteristics of impulse radio significantly improve the state of theart. Antennas employed in such impulse radio communication systems (aswell as in other impulse radio systems, such as positioning systems orother systems) have special requirements for effecting efficientoperation. Conflicting design parameters for design goals such asimproving uniformity of antenna propagation pattern improving impedancematching and improving antenna gain may be partially accommodated byemploying a novel signal coupling apparatus according to the presentinvention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

[0132] As mentioned earlier herein in the Summary of the Invention, anobject of the present invention to provide a small omni-directionalplanar dipole antenna that can radiate energy efficiently with minimalreflection and distortion. One attempt at providing such an antenna ispresented by U.S. Pat. No. 5,319,377 by Mike Thomas for “WidebandArrayable Planar Radiator”, issued Jun. 7, 1994 (hereinafter referred toas “Thomas”). Thomas discloses a circular planar dipole antenna for usewith a multi-band discrete frequency system. Thomas discloses that acircular shape for his antenna elements provides a propagation patternin which “the majority of generated waves propagate perpendicular to theplane of the antenna element” [Thomas; Col. 3, Lines 49-51]. Theinventor has discovered that this conclusion by Thomas is erroneous. Thepresent provides a structure that improves on Thomas' design in severalways.

[0133] The inventor has discovered that deviating from the circularshape of Thomas' design provides improved impedance matching propertiesto dipole antennas. In particular, within limits, greater eccentricityof radiating elements provides a better impedance match and reducesundesired reflections.

[0134] FIGS. 9(A) through (D) are schematic illustrations ofrepresentative planar dipole antennas with elliptical elements havingvarious ratios of major-to-minor axes. In FIGS. 9(A)-(D), a plurality ofdifferently shaped radiating elements in a variety of dipole antennadevices are illustrated on a common major axes 910, 912 and respectiveminor axes 914 a, 914 b, 914 c, 914 d. In particular, in FIG. 9(A), aplanar dipole antenna device 920 includes substantially similar, andpreferably substantially congruent radiating elements 922, 924.Radiating element 922 is oriented about major axis 910 and minor axis914 a. Radiating element 922 has a minor axis dimension m_(a) and amajor axis dimension M_(a). Radiating element 924 is oriented aboutmajor axis 912 and minor axis 914 a. Radiating element 924 has a minoraxis dimension m_(a) and a major axis dimension M_(a). Radiatingelements 922, 924 are separated by a gap G. For radiating elements 922,924, m_(a) is equal with M_(a); radiating elements 922, 924 are circularradiating elements similar to the antenna elements disclosed by Thomas.

[0135] In FIG. 9(B), a planar dipole antenna device 930 includessubstantially congruent radiating elements 932, 934. Radiating element932 is oriented about major axis 910 and minor axis 914 b. Radiatingelement 932 has a minor axis dimension m_(b) and a major axis dimensionM_(b). Radiating element 934 is oriented about major axis 912 and minoraxis 914 b. Radiating element 934 has a minor axis dimension m_(b) and amajor axis dimension M_(b). Radiating elements 932, 934 are separated bya gap G. For radiating elements 932, 934, m_(b) is related to M_(b)according to the following relation:

[0136] M_(b):m_(b) as 1.25:1

[0137] Thus, radiating elements 932, 934 are elliptical radiatingelements having a greater dimension along axes 910, 912 than thedimension along axis 914 b.

[0138] In FIG. 9(C), a planar dipole antenna device 940 includessubstantially congruent radiating elements 942, 944. Radiating element942 is oriented about major axis 910 and minor axis 914 c. Radiatingelement 942 has a minor axis dimension m_(c) and a major axis dimensionM_(c). Radiating element 944 is oriented about major axis 912 and minoraxis 914 c. Radiating element 944 has a minor axis dimension m_(c) and amajor axis dimension M_(c). Radiating elements 942, 944 are separated bya gap G. For radiating elements 942, 944, m_(c) is related to M_(c)according to the following relation:

[0139] M_(c):m_(c) as 1.50:1

[0140] Thus, radiating elements 942, 944 are elliptical radiatingelements having a greater dimension along axes 910, 912 than thedimension along axis 914 c.

[0141] In FIG. 9(D), a planar dipole antenna device 950 includessubstantially congruent radiating elements 952, 954. Radiating element952 is oriented about major axis 910 and minor axis 914 d. Radiatingelement 952 has a minor axis dimension m_(d) and a major axis dimensionM_(d). Radiating element 954 is oriented about major axis 912 and minoraxis 914 d. Radiating element 954 has a minor axis dimension m_(d) and amajor axis dimension M_(d). Radiating elements 952, 954 are separated bya gap G. For radiating elements 952, 954, m_(d) is related to M_(d)according to the following relation:

[0142] M_(d):m_(d) as 1.75:1

[0143] Thus, radiating elements 952, 954 are elliptical radiatingelements having a greater dimension along axes 910, 912 than thedimension along axis 914 d.

[0144]FIG. 10 is a schematic diagram of detail of an antenna feedstructure for a spheroidal dipole antenna. In FIG. 10, a spheroidaldipole antenna 1000 includes a first spheroidal radiating element 1002,a second spheroidal radiating element 1004 and a feed structure 1006.Feed structure 1006 is illustrated as a coaxial feed structure includinga center conductor 1008 substantially surrounded by a sleeve 1010. Aspace 1012 between center conductor 1008 and sleeve 1010 may be occupiedby air or by a dielectric material. Illustrating feed structure 1006 asa coaxial feed structure is merely illustrative and is not intended tolimit the variety of transmission lines or connectors that could beemployed in constructing feed structure 1006. A feed structure that isoriented substantially about the axis of an antenna is generallypreferred because energy flow and surface currents are minimized at theaxial locus.

[0145] Feed structure 1006 is coupled with antenna 1000 in a feed region1015. Center conductor 1008 is connected with first radiating element1002 at a feed point 1014 within feed region 1015, and sleeve 1010 isconnected with second radiating element 1004 at a connection locus 1016within feed region 1015. If additional mechanical strength or improvedresistance to electrical breakdown is desired, dielectric material maybe included in feed region 1015.

[0146] Variation in overall spheroidal geometry of radiating elements1002, 1004 of antenna 1000 may be accommodated without significantlyaffecting the performance of antenna 1000. The inventor has learned thatfeed region 1015 is critical to provide good matching and minimalreflection while operating antenna 1000. Prior art teaching hasgenerally asserted that a region at which an antenna is connected withits feed should be point-like at a feed point and flare out from thatfeed point. The present invention incorporates an antenna feed regionhaving a “blunt” or curved surface at a feed point, such as curvedsurface 1017 spanning a dimension “X₁” about feed point 1014, and curvedsurface 1019 spanning a dimension “X₂” about feed locus 1016 within feedregion 1015. The inventor has learned that it is advantageous to providean approximately spheroidal surface for connecting feed structure 1006.Such curved surfaces 1017, 1019 at an antenna feed point 1014 or anantenna feed locus 1016 significantly improve the impedance match of thejuncture between feed structure 1006 and radiating elements 1002, 1004within feed region 1015, thereby providing an improved match to 50 Ωthat is not so easily attainable using prior art antenna feedarrangements (if such a preferred low impedance is attainable at all).

[0147] A gap width “G” between radiating elements 1002, 1004 isestablished by the arrangement illustrated in FIG. 10. Gap width G is acritical parameter that must be carefully arranged for providing bestresults using antenna 1000. A gap width G approximately equal todiameter D of feed structure 1006 is a preferred starting dimension forbeginning adjustments to optimize performance.

[0148] In exemplary antenna 1000, feed structure 1006 embodies an energyguiding means, radiating elements 1002, 1004 cooperate to embody anenergy channeling structure and feed region 1015 embodies a transitionmeans.

[0149]FIG. 11(A) is a side view of a right angle coaxial connector feedstructure with a planar antenna. In FIG. 11(A), an antenna assembly 1100includes a dielectric substrate 1102 carrying a first radiating element1104 and a second radiating element 1106. A coaxial connector 1108provides a connection structure 1110 for a coaxial cable (not shown inFIG. 11(A)), and a right-angle structure 1112. Coaxial connector 1108 isaffixed with dielectric substrate 1102 incorporating spacer structure1114. Spacer structure 1114 may, for example, include a plurality ofnylon spacers, or another spacer structure appropriate to establish agap dimension “X” from dielectric substrate 1102 appropriate for properantenna operation by antenna assembly 1100. Ground pins 1116 (only oneground pin 1116 is visible in FIG. 11(A)) connect first radiatingelement 1104 with coaxial connector 1108. Center pin 1120 connectssecond radiating element 1106 with the center connector wire of thecoaxial cable (not shown in FIG. 11(A)) attached using coaxial connector1108.

[0150]FIG. 11(B) is a side view of a straight coaxial connector feedstructure with a planar antenna. In FIG. 11(B), an antenna assembly 1150includes a dielectric substrate 1152 carrying a first radiating element1154 and a second radiating element 1156. A coaxial connector 1158provides a connection structure 1160 for a coaxial cable (not shown inFIG. 11(B)). Coaxial connector 1158 is affixed with dielectric substrate1152 incorporating spacer structure 1164. Spacer structure 1164 may, forexample, include a plurality of nylon spacers, or another spacerstructure appropriate to establish a gap dimension “X” from dielectricsubstrate 1152 appropriate for proper antenna operation by antennaassembly 1150. Ground pins 1166 (only one ground pin 1166 is visible inFIG. 11(B)) connect first radiating element 1154 with coaxial connector1158. Center pin 1170 connects second radiating element 1156 with thecenter connector wire of the coaxial cable (not shown in FIG. 11(B))attached using coaxial connector 1158.

[0151]FIG. 11(C) is a top view of a curved feed interface arrangementfor an antenna of the sort illustrated in FIG. 11(A) or FIG. 11(B) takenalong Section 11C-11C of FIG. 11(A) or FIG. (B). In FIG. 11(C),radiating elements 1104, 1106 are carried upon dielectric substrate1102. In feed region 1115, ground pins 1116 are connected with firstradiating element 1104, and center pin 1120 is connected with secondradiating element 1106. Connection may be effected using solder or otherknown connection techniques. A gap G is established between radiatingelements 1104, 1106.

[0152]FIG. 12 is a schematic plan view of a planar antenna configuredaccording to the construction illustrated in FIG. 11. In FIG. 12, anantenna assembly 1200 includes a substantially planar substrate 1202 andradiating elements 1204, 1206 arrayed on substrate 1202. Radiatingelements 1204, 1206 are preferably substantially planar and made ofelectrically conductive material, such as copper. Radiating elements1202, 1204 are preferably adhered to or otherwise affixed upon substrate1202. Radiating elements are coupled with a signal supply 1208. Signalsupply 1208 is preferably a coaxial signal line of the sort representedin FIG. 11 (coaxial connectors 1108, 1158) and connected with radiatingelements 1204, 1206 in a manner of the sort described in detail in FIG.11. For some applications, it may be desirable for antenna assembly 1200to be conformal to a curved or other non-planar surface. Substrate 1202may be a flexible printed circuit board or other dielectric such as aglass window, fiberglass panel, plastic pipe, or any other suitabledielectric material.

[0153] The inventor has discovered that a more uniform pattern ofpropagation about the axis of the antenna elements (e.g., axis 914 a ofantenna radiating elements 922, 924; FIG. 9) is produced using lesseccentric radiating elements. In the case of circular elements, theradiation is essentially omni-directional. Thomas mistakenly teachesthat radiation from circular elements propagates principallyperpendicular to the plane of the radiating elements. The lesson fromexperiments by the inventor teach that directivity in a directionparallel to the plane containing the radiating elements is enhanced byeccentricity of the radiating elements. That is, a greater eccentricityof radiating elements produces greater directivity of propagationparallel to the plane containing the radiating elements. The result ofsuch greater directivity of propagation parallel to the plane containingthe radiating elements is that the entire propagation pattern for sucheccentric elements is less uniformly omni-directional and more likely toexhibit lobes of radiation intensity that disrupt uniformity of theradiation pattern. Conversely, lesser eccentricity (i.e., greatercircularity) of radiating elements produces improved uniformity ofpropagation of signals; i.e., more omni-directional propagation.

[0154] Moreover, greater eccentricity presents a broader, more gradualtransition from signal supply to antenna elements in the feed region sothat greater eccentricity yields better impedance matching, less signalreflection and an improves voltage standing wave ratio (VSWR).

[0155] The goal of producing an omni-directional antenna is incompatiblewith the goal of reducing reflection and improving impedance matching.Lesser eccentricity of antenna elements produces greater directionalpattern; increased eccentricity of antenna elements produces improvedimpedance matching and reduced signal reflection. Thus, there is adichotomous relation involved in antenna design as it pertains to dipoleantennas and eccentricity of radiating elements in such antennas. Thepresent invention provides greater freedom of design in connection withthis dichotomous relation.

[0156]FIG. 13 is a schematic plan view of a planar dipole antennaconfigured according to the preferred embodiment of the presentinvention. In FIG. 13, an antenna assembly 1300 includes a substantiallyplanar substrate 1302 and radiating elements 1304, 1306 arrayed onsubstrate 1302. Radiating elements 1304, 1306 are preferablysubstantially planar and made of electrically conductive material, suchas copper. Radiating elements 1304, 1306 are preferably adhered to orotherwise affixed upon substrate 1302. Radiating element 1304 is coupledwith a feed structure 1320. Preferably, radiating element 1304 and feedstructure 1320 are integrally formed in a single conductive piece. Feedstructure 1320 comprises a transition structure that preferably dividesradiating element 1306 into two sections 1306 a, 1306 b and establishesa separation 1322 between section 1306 a and feed structure 1320 and aseparation 1324 between section 1306 b and feed structure 1320.

[0157] A signal supply 1308 couples antenna assembly 1300 with a hostdevice (not shown in FIG. 13). Signal supply 1308 is preferably acoaxial connector of the sort represented in FIG. 11 (e.g., coaxialconnector 1158). Signal supply 1308 is connected with feed structure1320 via a center pin 1310. Signal supply 1308 is connected with section1306 a of radiating element 1306 via a ground pin 1312 and signal supply1308 is connected with section 1306 b of radiating element 1306 via aground pin 1313. Connecting center pin 1310 with feed structure 1320effects connection of center pin 1310 with radiating element 1304.

[0158] Preferably separations 1322, 1324 vary from a narrow expanseproximate signal supply 1308 to a relatively wider expanse distal fromsignal source 1308 and proximate to radiating element 1304. Thus,separations 1322, 1324 operate to vary impedance from a relatively lowervalue proximate with signal supply 1308 to a relatively higher valueproximate with radiating element 1304. It is in this varying of expanseof separations 1322, 1324 and the consequent variance of impedance thatfeed structure 1320 operates as a transition element in effectingcoupling between antenna assembly 1300 and signal supply 1308. Thus, afeed region 1316 is established over a feed distance from a firstconnection locus 1315 proximal with signal supply 1308 to a secondconnection locus 1317 proximal with radiating structure 1304. One mayalter the variance maximum and minimum of separations 1322, 1324, or onemay alter the rate of change of the variance over the feed distance, orone may effect both such alterations in adjusting the impedance of thecoupling between antenna assembly 1300 and signal supply 1308.

[0159] It is this capability to directly affect impedance matching bymaking such alterations in expanses of separations 1322, 1324 thatallows more independent adjustment of eccentricity of radiating elements1304, 1306 in designing antenna assembly 1300. The improved structure ofthe present invention enables one to directly affect impedance matchingindependently of adjusting eccentricity of radiating elements 1304,1306. This difference in design flexibility allows one to require lesseccentricity to accomplish a given impedance matching than was requiredusing the prior art design (e.g., FIGS. 11, 12). This is so because atleast a portion of the adjustment for improved impedance matching may beaccomplished by adjusting expanses of separations 1322, 1324independently of adjusting eccentricity of radiating elements 1304,1306.

[0160]FIG. 14 is a schematic perspective view of a spheroidal dipoleantenna configured according to the present invention. In FIG. 14, anantenna assembly 1400 three-dimensional radiating elements 1404, 1406.Preferably radiating elements 1404, 1406 are spheroidal or ellipsoidalin shape, coaxially situated on an axis 1405 and made of electricallyconductive material, such as copper. Radiating element 1404 is coupledwith a feed structure 1420. Preferably, radiating element 1404 and feedstructure 1420 are integrally formed in a single conductive piece. Feedstructure 1420 comprises a transition structure that preferablytraverses radiating element 1406 along axis 1405 and establishes aseparation 13422 between section 1406 and feed structure 1420.

[0161] A signal supply 1408 couples antenna assembly 1400 with a hostdevice (not shown in FIG. 14). Signal supply 1408 is preferably acoaxial signal line of the sort represented in FIG. 11 (e.g., coaxialconnector 1158). Signal supply 1408 is connected with feed structure1420 via a center pin 1410. Signal supply 1408 is connected withradiating element 1406 via at least one of a ground pin 1412 and aground pin 1413. Connecting center pin 1410 with feed structure 1420effects connection of center pin 1410 with radiating element 1404.

[0162] Preferably separation 1422 varies from a narrow expanse proximatesignal supply 1408 to a relatively wider expanse distal from signalsource 1408 and proximate to radiating element 1404. Thus, separation1422 operates to vary impedance from a relatively lower value proximatewith signal supply 1408 to a relatively higher value proximate withradiating element 1404. It is in this varying of expanse of separation1422 and the consequent variance of impedance that feed structure 1420operates as a transition element in effecting coupling between antennaassembly 1400 and signal supply 1408. Thus, a feed region 1416 isestablished over a feed distance from a first connection locus 1415proximal with signal supply 1408 to a second connection locus 1417proximal with radiating structure 1404. One may alter the variancemaximum and minimum of separation 1422, or one may alter the rate ofchange of the variance over the feed distance, or one may effect bothsuch alterations in adjusting the impedance of the coupling betweenantenna assembly 1400 and signal supply 1408.

[0163] It is this capability to directly affect impedance matching bymaking such alterations in expanse of separation 1422 that allows moreindependent adjustment of eccentricity of radiating elements 1404, 1406in designing antenna assembly 1400. The improved structure of thepresent invention enables one to directly affect impedance matchingindependently of adjusting eccentricity of radiating elements 1404,1406. This difference in design flexibility allows one to require lesseccentricity to accomplish a given impedance matching than was requiredusing the prior art design (e.g., FIGS. 11, 12 and three-dimensionalversions of such prior art antenna design). This is so because at leasta portion of the adjustment for improved impedance matching may beaccomplished by adjusting expanse of separation 1422 independently ofadjusting eccentricity of radiating elements 1404, 1406. A variety ofsuitable radiating elements may be generated, for example, from surfacesof revolution that may be obtained by rotating the characteristic planarelements of FIG. 9 about their semi-minor axes 914 a-d. In a preferredembodiment, radiating elements 1404 and 1406 are ellipsoids ofrevolution whose semi-major and semi-minor axes are in a ratio ofapproximately 1.5:1. This yields a shape which is approximatelyconformal to the natural contours of energy flow around an ideal dipole.Further, such a shape significantly reduces undesired reactive energyand gives rise to the desired ultra-wideband response and excellentimpedance matching.

[0164] Antennas with feed regions similar to feed region 1416 have beenused in the past, as disclosed for example by Brillouin in U.S. Pat. No.2,454,766 (issued Nov. 30, 1948). Brillouin used a tapered coaxial feedwith a coaxial horn antenna having a propagation path extending to theedges, or rims of the horn structures of his antenna. Brillouin did notteach or suggest using a tapered coaxial feed structure with a dipoleantenna as shown in FIG. 14.

[0165] There is also a third significant consideration that must betaken into account in designing ultra-wide band planar dipole antennas:current induced in the signal supply, such as sheath current induced ona coaxial signal supply line. Sheath currents occur whenever a dipoleantenna (an inherently balanced structure) is fed by a coaxial line (aninherently unbalanced structure) or a similarly unbalanced feed line.Sheath currents may also be caused when feed lines are improperly routedthrough regions in which radiation fields pass. Sheath currentsgenerally make it difficult to achieve better than about 10 dB isolationbetween the operating band of an antenna and the lower frequency “stopband” in which the antenna is not intended to operate. Because antennacables carrying sheath currents are radiators of energy, overall systemperformance becomes difficult to predict, and undesired out-of-bandradiation may occur.

[0166] A further benefit is realized by the structure of the presentinvention in that the ability to control antenna impedance to a highdegree of precision allows accurate design control of antenna gain rolloff at the edge of the pass band of the antenna. Varying the impedancetaper of the transformation structure of the present invention allows adesigner to vary the rate of roll-off from a gentle slope (e.g., about−10 dB per octave) to a steeper slope (e.g., about −20 dB per octave).Thus, the present invention's provision of a variable impedance inputcoupling to an antenna facilitates use of the antenna itself as abroadband filter device with a variable roll-off characteristic.

[0167] A still further benefit of the present invention is therelatively small size of the antenna. The minor axis has a size that isroughly 0.125 wavelength at the lower end of the operating band. This isabout half the size of the usual quarter wave structure previouslybelieved in the art to be necessary for efficient performance.

[0168] In addition, if a narrow band implementation of the presentinvention is desired, there are a variety of methods known in the art toreduce the upper operating frequency while maintaining the same loweroperating frequency. These include but are not necessarily limited tointroducing a frequency dependent loss, or another means for effectinglow pass filtering. This would allow an antenna designer to use thepresent teachings to create very small, very efficient narrow bandantennas.

[0169]FIG. 15 is a graphic representation of a typical relationshipbetween antenna gain and signal frequency for antennas configuredaccording to the teachings of the present invention, includingadjustment of gain roll-off for such antennas. In FIG. 15, a graphicchart 1500 plots antenna gain for a representative ultra-wide band (UWB)antenna on an axis 1502 versus operating frequency of the antenna on anaxis 1504. Thus, gain response as a function of frequency is representedby a response curve 1510 on chart 1500. Values displayed on axes 1502,1504 are representative and are not intended to limit the scope of thedisclosure. For example, an antenna designer can obtain an antennaaccording to the teachings of the present invention in a differentdesired operating band by scaling the physical dimensions of the antennaappropriately.

[0170] Response curve 1510 extends from a lower-frequency region 1512 toan upper-frequency region 1518 to establish an operating band “A” and anoperating band “B”. Operating band “A” extends from about 2 GHz to about6 GHz and represents a normal UWB dipole operating band. Operating band“B” extends upward in frequency from about 6 GHz and represents anoperating band in which quadrupole operation may be experienced. Thewide swings in gain as a function of frequency in lower region 1512represent variations in gain caused by poor isolation in a signalsupply, such as sheath currents in a coaxial cable signal supply. Dottedline curve 1522 represents a relatively gentle gain roll-off of about−10 dB per octave from operating band “A” for one geometry of feedregion (e.g., feed region 1316, FIG. 13; or 1416, FIG. 14) with a givenseparation (e.g., separations 1322, 1324, FIG. 13; separation 1422, FIG.14) between radiating element(s) and a feed structure (e.g., feedstructure 1320, FIG. 13; feed structure 1420, FIG. 14).

[0171] Dotted line curve 1524 represents a relatively sharper gainroll-off of about −20 dB per octave from operating band “A” for anothergeometry of feed region (e.g., feed region 1316, FIG. 13; or 1416, FIG.14) with another given separation (e.g., separations 1322, 1324, FIG.13; separation 1422, FIG. 14) between radiating element(s) and a feedstructure (e.g., feed structure 1320, FIG. 13; feed structure 1420, FIG.14). Such a capability to adjust gain roll-off enables an antennadesigner to avoid noisy areas such as the wide swings in gain in lowerregion 1512 caused by poor isolation in a signal supply, such as sheathcurrents in a coaxial cable signal supply, as illustrated by dotted linecurve 1524.

[0172]FIG. 16 is a schematic plan view of a planar dipole antennaconfigured according to an alternate embodiment of the presentinvention. The alternate embodiment of the present invention illustratedin FIG. 16 includes a balun transformer structure in the antenna feedregion. A balun is a special kind of transformer that transforms anunbalanced transmission line to a balanced transmission line. If anunbalanced line is used to drive a balanced antenna, the resulting feedasymmetry leads to undesired currents and reflection back down thetransmission line. A balun allows an unbalanced feed structure to becoupled with a balanced antenna without such unwanted effects. Such abalun embodiment is useful, for example, when sheath currents are likelyto cause problems or when good stop band protection is desired. In FIG.16, a first side 1602 of an antenna assembly 1600 is illustrated in FIG.16(A), and a second side 1604 of antenna assembly 1600 is illustrated inFIG. 16(B). Second side 1604 is opposite to first side 1602.

[0173] Antenna assembly 1600 includes a substantially planar substrate1605 with radiating elements 1606, 1608 arrayed on substrate 1605.Radiating elements 1606, 1608 are preferably substantially planar andmade of electrically conductive material, such as copper. Radiatingelement 1606 is preferably adhered to or otherwise affixed upon firstside 1602 of substrate 1605. Radiating element 1608 is preferablyadhered to or otherwise affixed upon second side 1604 of substrate 1605.

[0174] Radiating element 1606 is coupled with a first feed structure1620. Preferably, radiating element 1606 and first feed structure 1620are integrally formed in a single conductive piece on first side 1602 ofantenna assembly 1600. First feed structure 1620 divides radiatingelement 1606 into two sections 1606 a, 1606 b and establishes aseparation 1622 between section 1606 a and first feed structure 1620 anda separation 1624 between section 1606 b and first feed structure 1620.An isolation gap G1 is established between first feed structure 1620 andsection 1606 a, and an isolation gap G2 is established between firstfeed structure 1620 and section 1606 b. Isolation gaps G1, G2 must besufficiently wide to preclude undesired coupling among first feedstructure 1620 and sections 1606 a, 1606 b.

[0175] Radiating element 1608 is coupled with a second feed structure1630. Preferably, radiating element 1608 and second feed structure 1630are integrally formed in a single conductive piece on second side 1604of antenna assembly 1600. Second feed structure 1630 is preferablysubstantially symmetrically situated on second side 1604 with respect tofirst feed structure 1620 on first side 1602.

[0176] Signal supply 1609 couples antenna assembly 1600 with a hostdevice (not shown in FIG. 16). Signal supply 1609 is preferably acoaxial connector of the sort represented in FIG. 11 (e.g., coaxialconnector 1158). Signal supply 1609 is connected with first feedstructure 1620 via a center pin 1610. Signal supply 1609 is connectedwith second feed structure 1630 via ground pins 1612, 1613. Connectingcenter pin 1610 with first feed structure 1620 effects connection ofcenter pin 1610 with radiating element 1606 because sections 1606 a,1606 b are coupled with first feed structure 1620 on first side 1602 atconnection locus 1640. Connecting ground pins 1612, 1613 with secondfeed structure 1630 effects connection of ground pins 1612, 1613 withradiating element 1608 because radiating element 1608 is coupled withsecond feed structure 1630 on second side 1604 at connection locus 1642.Thus, feed structures 1620, 1630 cooperate to establish a feed region1644 for antenna assembly 1600.

[0177] Second feed structure 1630 has a wider expanse than first feedstructure 1620 at a locus proximate with a signal supply 1609. Secondfeed structure 1630 converges to a narrower width at coupling locus1642. First feed structure 1620 has a narrower expanse than second feedstructure 1630 at a locus proximate with a signal supply 1609. Firstfeed structure 1620 diverges to a wider width at coupling locus 1640.The convergence-divergence of widths of feed structures 1620, 1630provide a capability for balancing signals applied at signal supply1609. Widths of feed structures 1620, 1630 at their respectiveconnection loci 1640, 1642 affect coupling impedance of antenna assembly1600.

[0178] Thus, antenna assembly 1600 provides substantially independentadjustment of signal balance (i.e., by adjusting convergence-divergenceof feed structures 1620, 1630) and adjustment of coupling impedance(i.e., by adjusting widths of joining between feed structures 1620, 1630and radiating elements 1606, 1608 at connection loci 1640, 1642).

[0179] It is this capability to substantially independently affectimpedance matching and signal balance that allows more independentadjustment of eccentricity of radiating elements 1606, 1608 in designingantenna assembly 1600. The improved structure of the present inventionenables one to substantially independently affect impedance matching,signal balance and eccentricity of radiating elements 1606, 1608. Thisdifference in design flexibility allows one to require less eccentricityto accomplish a given impedance matching while still efficientlycoupling an unbalanced signal feed with a balanced antenna than wasrequired using the prior art design (e.g., FIGS. 11, 12). This is sobecause at least a portion of the adjustment for improved impedancematching and balanced coupling may be accomplished by adjustingconvergence-divergence of feed structures 1620, 1630, and by adjustingwidths of joining between feed structures 1620, 1630 and radiatingelements 1606, 1608 at connection loci 1640, 1642.

[0180] Thus, in summary, there are three principal significant designconsiderations that should be taken into account in designing anultra-wide band (UWB) antenna: (1) if an omni-directional propagationpattern is desired, antenna radiation elements should be less eccentric(i.e., more circular) in shape; (2) if good impedance matching isdesired, antenna radiation elements should be more eccentric (i.e., lesscircular) in shape; and (3) if reliable performance and good stop bandisolation are desired, feed line sheath currents must be kept small.

[0181] Prior art antenna designs have largely comprised attaching asignal supply (e.g., a coaxial cable) to a slot line between two antennaradiating elements and living with whatever performance is provided bythe arrangement. Eccentricity of radiating elements could be increasedto improve impedance matching, but only at the expense of uniformity ofpropagation pattern. Thus, a designer had to strive to “strike abalance” between desired impedance matching and uniformity ofpropagation. Shortcomings in either parameter had to be made up usingtechniques external to the antenna structure. Such external techniquesincluded, by way of example, resistively loading the input impedance toimprove impedance matching or providing selected reflection structuresto reshape propagation patterns. Resistive loading adversely affectsantenna efficiency; reflection can help aim propagation in a givendirection, but it is not a solution for providing omni-directionalpropagation.

[0182] The present invention provides a structure for coupling a signalsupply (such as a coaxial line) with antenna radiation elements thatpermits a more forgiving transformation in the feed region of theantenna structure than is established by prior art structures. Byestablishing such an impedance transformation region a designer isprovided with a structure that can be varied to more directly affectimpedance matching than can be effected merely by varying eccentricityof radiating elements. The result is that, while eccentricity stillaffects uniformity of propagation and impedance matching (as it does inthe case of earlier antenna designs), there is an additional adjustmentprovided by the present invention that more directly affects impedancematching. As a consequence, eccentricity may now be skewed closer to thecircular shape desired to facilitate uniformity of propagation than waspreviously achievable, while still attaining acceptable impedancematching for an antenna design. What occurs, in effect, is a “pulling”into the feed region of the gradual transition exhibited by eccentricelliptical radiating elements. As a happy consequence, the betterimpedance matching provided by a more eccentric elliptical radiatingelement can now be realized while employing a less eccentric morecircular radiating element to effect acceptable uniformity in signalpropagation.

[0183] Yet a further benefit of the present invention is that cablecurrent can be significantly reduced by routing the feed supply of thepresent invention along the axis of the antenna. Such an orientation ofthe feed supply avoids placing the feed supply in the direct path ofradiated fields, thus reducing the incidence of sheath currents in thefeed supply.

[0184] It is to be understood that, while the detailed drawings andspecific examples given describe preferred embodiments of the invention,they are for the purpose of illustration only, that the apparatus andmethod of the invention are not limited to the precise details andconditions disclosed and that various changes may be made thereinwithout departing from the spirit of the invention which is defined bythe following claims:

I claim:
 1. An apparatus for coupling a balanced antenna with a signalline; said signal line having at least two signal line coupling loci fordelivering a signal having at least two unbalanced signal components;said antenna having at least two antenna coupling loci for receivingsignals; the apparatus comprising: (a) a first feed element; said firstfeed element being coupled with a first signal line coupling locus ofsaid at least two signal line coupling loci; said first feed elementconveying a first signal component of said at least two signalcomponents along a first feed element length to a first antenna couplinglocus of said at least two antenna coupling loci; and (b) a second feedelement; said second feed element being coupled with a second signalline coupling locus of said at least two signal line coupling loci; saidsecond feed element conveying a second signal component of said at leasttwo signal components along a second feed element length to a secondantenna coupling locus of said at least two antenna coupling loci; saidfirst signal component and said second signal component differing in atleast one parameter by an imbalance; said first feed element and saidsecond feed element being separated by a dielectric gap; said first feedelement and said second feed element being mutually asymmetric; saidmutual asymmetry being configured to reduce said imbalance as said firstsignal component and said second signal component traverse the apparatusintermediate said signal line and said antenna.
 2. An apparatus forcoupling a balanced antenna with a signal line as recited in claim 1wherein said first feed element and said second feed element are eachsubstantially planar quadrilateral lands; said first feed element havinga first signal coupling width neighboring said first signal linecoupling locus and having a first antenna coupling width neighboringsaid first antenna coupling locus; said second feed element having asecond signal coupling width neighboring said second signal linecoupling locus and having a second antenna coupling width neighboringsaid second antenna coupling locus; said first signal coupling width andsaid second signal coupling width being unequal by a signal couplingwidth difference.
 3. An apparatus for coupling a balanced antenna with asignal line as recited in claim 1 wherein said first feed element andsaid second feed element are each substantially planar quadrilaterallands; said first feed element having a first signal coupling widthneighboring said first signal line coupling locus and having a firstantenna coupling width neighboring said first antenna coupling locus;said second feed element having a second signal coupling widthneighboring said second signal line coupling locus and having a secondantenna coupling width neighboring said second antenna coupling locus;said first antenna coupling width and said second antenna coupling widthbeing unequal by an antenna coupling width difference.
 4. An apparatusfor coupling a balanced antenna with a signal line as recited in claim 1wherein said first feed element and said second feed element are eachsubstantially planar quadrilateral lands; said first feed element havinga first signal coupling width neighboring said first signal linecoupling locus and having a first antenna coupling width neighboringsaid first antenna coupling locus; said second feed element having asecond signal coupling width neighboring said second signal linecoupling locus and having a second antenna coupling width neighboringsaid second antenna coupling locus; said first signal coupling width andsaid second signal coupling width being unequal by a signal couplingwidth difference; said first antenna coupling width and said secondantenna coupling width being unequal by an antenna coupling widthdifference.
 5. An apparatus for coupling a balanced antenna with asignal line as recited in claims 2 through 4 wherein said first feedelement and said second feed element are substantially coplanar andwherein said dielectric gap is an air gap intermediate said first feedelement and said second feed element.
 6. An apparatus for coupling abalanced antenna with a signal line as recited in claims 2 through 4wherein said first feed element is situated on a first side of adielectric substrate and wherein said second feed element is situated ona second side of said dielectric substrate; said dielectric substratehaving a thickness separating said first side and said second side; saidthickness establishing said dielectric gap.
 7. An apparatus for couplinga balanced antenna with a signal line as recited in claim 2 wherein saidmutual asymmetry is configured to reduce said imbalance by said signalcoupling width difference.
 8. An apparatus for coupling a balancedantenna with a signal line as recited in claim 3 wherein said mutualasymmetry is configured to reduce said imbalance by said antennacoupling width difference.
 9. An apparatus for coupling a balancedantenna with a signal line as recited in claim 4 wherein said mutualasymmetry is configured to reduce said imbalance by at least one of saidsignal coupling width difference and said antenna coupling widthdifference.
 10. An apparatus for establishing signal coupling between asignal line and an antenna structure; said antenna structure including afirst radiating element on a first side of a dielectric substrate and asecond radiating element on a second side of said substrate; said firstradiating element having a first antenna coupling locus; said secondradiating element having a second antenna coupling locus; said signalline having two signal line coupling loci for delivering a signal havingtwo unbalanced signal components; the apparatus comprising: (a) a firstfeed element on said first side of said substrate; said first feedelement being a first substantially planar polygonal land and couplingsaid first antenna coupling locus with a first signal line couplinglocus of said two signal line coupling loci; said first feed elementestablishing a first antenna coupling width at said first antennacoupling locus and establishing a first signal coupling width at saidfirst signal line coupling locus; and (b) a second feed element on saidsecond side of said substrate; said second feed element being a secondsubstantially planar polygonal land and coupling said second antennacoupling locus with a second signal line coupling locus of said twosignal line coupling loci; said second feed element establishing asecond antenna coupling width at said second antenna coupling locus andestablishing a second signal coupling width at said second signal linecoupling locus; said first antenna coupling width, said second antennacoupling width, said first signal coupling width and said second signalcoupling width cooperating to establish an asymmetry between said firstfeed element and said second feed element; said asymmetry operating toreduce imbalance between said two unbalanced signal components.
 11. Anapparatus for establishing signal coupling between a signal line and anantenna structure as recited in claim 10 wherein said asymmetry operatesto modulate said two unbalanced signal components to present balancedsignal components at said first antenna coupling locus and said secondantenna coupling locus.
 12. An apparatus for establishing signalcoupling between a signal line and an antenna structure as recited inclaim 10 wherein said substrate is substantially planar.
 13. Anapparatus for establishing signal coupling between a signal line and anantenna structure as recited in claim 11 wherein said substrate issubstantially planar.
 14. An apparatus for establishing signal couplingbetween a signal line and an antenna structure as recited in claim 12wherein said substrate is substantially inflexible.
 15. An apparatus forestablishing signal coupling between a signal line and an antennastructure as recited in claim 13 wherein said substrate is substantiallyinflexible.
 16. An apparatus for establishing signal coupling between anunbalanced signal line and an antenna structure; said antenna structureincluding a first radiating element and a second radiating element; saidunbalanced signal line including a first signal carrying elementcarrying a first signal component and a second signal carrying elementcarrying a second signal component; said first signal component and saidsecond signal component being unequal and establishing an unbalancedsignal; said first signal carrying element and said second signalcarrying element cooperating to deliver said unbalanced signal to saidantenna structure in an installed orientation at a connection locus;said first radiating element having at least one first proximal edge andat least one first distal edge with respect to said signal line in saidinstalled orientation; said second radiating element having at least onesecond proximal edge and at least one second distal edge with respect tosaid signal line in said installed orientation; said connection locusgenerally including a portion of said at least one first proximal edgeand a portion of said at least one second proximal edge; the apparatuscomprising: (a) a first feed structure; said first feed structureextending a feed distance from said signal line in said installedorientation to said second proximal edge; said first feed structuresubstantially dividing said first radiating element into at least twolands; said at least two lands being electrically common in spacedrelation with said first feed structure to establish a first separationdistance intermediate said first feed structure and at least a firstselected land of said at least two lands on a first side of said firstfeed structure substantially along said feed distance and to establish asecond separation distance intermediate said first feed structure and atleast a second selected land of said at least two lands on a second sideof said first feed structure; and (b) a second feed structure couplingsaid signal line with said first proximal edge; said first separationdistance and said second separation distance being dimensionedappropriately to establish a signal transmission structure between saidat least two lands and said first feed structure; said first signalcomponent and said second signal component differing in at least oneparameter by an imbalance; said first feed element and said second feedelement being mutually asymmetric; said mutual asymmetry beingconfigured to reduce said imbalance as said first signal component andsaid second signal component traverse the apparatus intermediate saidsignal line and said antenna structure.
 17. An apparatus forestablishing signal coupling between a signal line and an antennastructure as recited in claim 16 wherein said separation distance isvaried along said feed distance appropriately to establish a desiredimpedance in said signal coupling.
 18. An apparatus for establishingsignal coupling between a signal line and an antenna structure asrecited in claim 16 wherein said first radiating element, said secondradiating element and said first feed structure are substantially planarand are arrayed upon a substantially planar substrate.
 19. An apparatusfor establishing signal coupling between a signal line and an antennastructure as recited in claim 18 wherein said first radiating element,said second radiating element and said first feed structure are arrayedon a single side of said substrate.
 20. An apparatus for establishingsignal coupling between a signal line and an antenna structure asrecited in claim 17 wherein said first radiating element, said secondradiating element and said first feed structure are substantially planarand are arrayed upon a substantially planar substrate.
 21. An apparatusfor establishing signal coupling between a signal line and an antennastructure as recited in claim 20 wherein said first radiating element,said second radiating element and said first feed structure are arrayedon a single side of said substrate.